In the development of radio communication systems of the third (3G) and fourth (4G) generation, particular emphasis is put on the exploitation of the spatial domain of the transmission medium in order to increase the spectral efficiency (measured in bit/s/Hz) of these systems. To exploit the spatial domain, at least two antenna elements have to be provided at a transmitter and/or receiver of a radio communication system, for instance in the shape of an antenna array with at least two antenna elements. By base-band manipulation of the signals fed to these antenna elements or received from these antenna elements, it becomes then possible to transmit and/or receive signals in a spatially selective fashion, for instance to place a peak of the transmission and/or reception antenna characteristic of the antenna array towards the direction of a desired user and to place a transmission and/or reception null towards the direction of an interferer. The deployment of an antenna array in a radio communication system that uses Time Division, Frequency Division or Code Division Multiple Access (TDMA, FDMA, CDMA) to control the access of several users to the shared transmission medium further allows for the incorporation of a Space Division Multiple Access (SDMA) component. For instance, in a pure TDMA system, wherein users are separated in the time domain, it becomes possible to assign two or more users the same time slot for transmission, because their signals can be spatially separated under exploitation of the spatial domain. Similarly, if only one multi-antenna transmitter and one multi-antenna receiver is present, it becomes possible to establish several orthogonal spatial channels between said transmitter and receiver for the transmission of several concurrent data streams between the two of them.
The basic transmission and reception of signals in a radio communication system with multiple antennas at the transmit and/or receive site is described by the standard base-band space-time system model, assuming ideal D/A conversion and modulation at the transmit site and ideal demodulation and A/D conversion at the receive site:x(k)=G(k)z(k)+I(k).  (1)
Therein, k is a burst index, i.e. it is assumed that signals are transmitted in bursts containing a number of N symbols each. It is assumed that at a transmit site, MT transmit antenna elements are available, and that at a receive site, MR receive antenna elements are available. These transmit and receive antenna elements can be assigned to one transmitter or receiver, respectively, or be assigned to more than one transmitter or receiver. For instance, there may be one transmitter with an antenna array with MT antenna elements or MT single-antenna transmitters, and one receiver with an antenna array with MR antenna elements or MR single-antenna receivers. From each transmit antenna element mT=1, . . . , MT, one transmit signal with n=1, . . . , N symbols zmT(n) is transmitted. These MT transmit signals are stacked in the transmit signal vectorz(k)=[z1(1) . . . z1(N) . . . zMT(1) . . . zMT(N)]T.
Correspondingly, the symbols xmR(n) received at receive antenna element mR=1, . . . , MR at time instant n=1, . . . , N+v in response to the transmission of said transmit symbols zmT(n) are stacked in the receive signal vectorx(k)=[x1(1) . . . x1(N+v) . . . xMR(1) . . . xMR(N+v)]T,wherein the parameter v will be explained below.
The actual mapping of said transmit signal vector z(k) onto said receive signal vector x(k) is performed by the space-time Multiple Input Multiple Output (MIMO) channel matrix
                              G          ⁡                      (            k            )                          =                              (                                                                                                      G                                              1                        ,                        1                                                              ⁡                                          (                      k                      )                                                                                        ⋯                                                                                            G                                              1                        ,                                                  M                          T                                                                                      ⁡                                          (                      k                      )                                                                                                                    ⋮                                                  ⋱                                                  ⋮                                                                                                                        G                                                                        M                          R                                                ,                        1                                                              ⁡                                          (                      k                      )                                                                                        ⋯                                                                                            G                                                                        M                          R                                                ,                                                  M                          T                                                                                      ⁡                                          (                      k                      )                                                                                            )                    ∈                                    C                                                (                                      N                    +                    v                                    )                                ⁢                                                      M                    R                                    ⨯                                      NM                    T                                                                        .                                              (        2        )            
Therein, the space-time Single Input Single Output (SISO) blocks GmR,mT(k) are Toeplitz matrices that are composed of shifted versions of the channel impulse response of the physical transmission (or propagation) channel between transmit antenna mT and receive antenna mR with an assumed number of (v+1) taps. It is understood that the space-time MIMO channel matrix is assumed to be substantially time-invariant (i.e. constant) during the transmission of at least one burst. The space-time MIMO channel matrix contains all physical propagation effects between the MT transmit antenna elements and the MR receive antenna elements such as for instance propagation delay; path loss; large-scale fading due to shadowing; small-scale fading due to multi-path propagation; receivers or scatterers in the transmission channel; and scattering, diffraction and refraction effects. Furthermore, all hardware characteristics effecting the signal during the transmission from the transmitter baseband to the receiver baseband are lumped to the space-time MIMO channel matrix as well, as for instance pulse shape filters, correlations due to antenna coupling or calibration errors, phase shifts due to non-ideal mixing or lack of transmitter-receiver clock synchronism, or delays due to filtering times.
Finally, the vectorI(k)=[I1(1) . . . I1(N+v) . . . IMR(1) . . . IMR(N+v)]T contains the noise ImR(n) received at each time instant n=1, . . . , N+v at receive antenna element mR=1, . . . , MR. Said noise contribution may represent thermal noise as well as signals from interferers (for instance intra-cell, inter-cell and inter-system interferers).
If the transmission channel between at least one transmit antenna element mT and one receive antenna element mR is frequency-selective, i.e. the coherence bandwidth of said transmission channel is substantially smaller than the bandwidth of transmit signals that are to be transmitted over said transmission channel, even in the simple case of MT=1 transmit antenna and MR=1 receive antenna, the receive signal x(k) will be heavily distorted even in the noise-free case. To obtain a reconstruction of the transmit signal, these distortions have to be mitigated or removed. This may be accomplished before the transmission of said transmit signal, after the transmission, or before and after said transmission by equalization of the MIMO channel. Throughout this specification, the term equalization will be used for any signal processing at the transmitter and/or receiver site that at least partially aims at a mitigation or removal of signal distortions experienced during the transmission of signals over a transmission channel and/or at least partially aims at a mitigation or removal of interference between signals transmitted over said transmission channel and/or at least partially aims at an improvement of the Signal-to-Noise Ratio (SNR) or the Signal-to-Noise-and-Interference Ratio (SNIR) of at least one reconstruction or estimate of said signals transmitted over said transmission channel.
Time domain based equalization of the space-time MIMO channel generally involves inversions of the space-time MIMO channel matrix G(k) or functions thereof. For instance, if Zero-Forcing (ZF) equalization of the MIMO channel matrix at the receiver site is desired, the left pseudo-inverse of the space-time MIMO channel matrix G(k) has to be computed, which requires complex multiplications of the order (NMT)3. It is thus evident that with increasing numbers of transmitted symbols N and increasing numbers of transmit antenna elements MT, time domain based equalization of the MIMO channel becomes unfeasible.
A particular reduction of the computational complexity encountered in the equalization of the space-time MIMO channel can be achieved by exploiting the block-Toeplitz structure of the space-time MIMO channel matrix G(k). This is achieved by transforming the space-time SISO blocks GmR,mT(k) in the space-time MIMO channel matrix G(k) into circulant blocks {tilde over (G)}mR,mT(k) of dimension N×N (i.e., the i-th row is equal to the j-th row cyclically shifted by i-j elements), yielding a space-time MIMO channel matrix G(k) composed of circulant blocks. Note that, for ease of presentation, in the following part of the description, the burst index k will be skipped.
As shown in U.S. Pat. No. 6,144,711, cyclic matrices can be diagonalized via Fourier and inverse Fourier transformations. In the present case, the cyclic SISO blocks {tilde over (G)}mR,mT within {tilde over (G)} can be transformed into diagonal matrices DmR,mT by the following operation:
                                                        Y                              (                                  M                  R                                )                                      ⁢                          G              ~                        ⁢                          Y                              (                                  M                  T                                )                            H                                =                      (                                                                                D                                          1                      ,                      1                                                                                        ⋯                                                                      D                                          1                      ,                                              M                        T                                                                                                                                          ⋮                                                  ⋱                                                  ⋮                                                                                                  D                                                                  M                        R                                            ,                      1                                                                                        ⋯                                                                      D                                                                  M                        R                                            ,                                              M                        T                                                                                                                  )                          ,                            (        6        )            wherein
                              Y                      (            M            )                          =                  (                                                    Y                                                                                                                          0                                                                                                                                                  ⋱                                                                                                                                                  0                                                                                                                          Y                                              )                                    (        7        )            contains M N×N Fourier transformation matrices Y on its diagonal, the elements of which are defined as
            y              m        ,        n              =                  1                  N                    ⁢              ⅇ                              -            j2π                    ⁢                                          ⁢                      mn            /            N                                ,and wherein Y(M)H is the corresponding inverse Fourier transformation matrix. Note that the multiplication with the Fourier transformation matrix and with the inverse Fourier transformation matrix may be efficiently implemented via a Fast Fourier Transformation (FFT) and an Inverse Fast Fourier Transformation (IFFT), in particular when N is a power of 2. A block-diagonalization (the resulting matrix then has only blocks on the diagonal and zeros elsewhere) of the space-time MIMO channel matrix {tilde over (G)} can then be achieved by applying unitary permutation matrices PR and PT in the following way:
                                                                        P                R                            ⁢                              Y                                  (                                      M                    R                                    )                                            ⁢                              G                ~                            ⁢                              Y                                  (                                      M                    T                                    )                                H                            ⁢                              P                T                                      ≡            H                    =                      (                                                                                H                    ⁡                                          (                      1                      )                                                                                                                                                                                0                                                                                                                                                                      ⋱                                                                                                                                                                      0                                                                                                                                                              H                    ⁡                                          (                      N                      )                                                                                            )                          ,                            (        8        )            wherein H is a block-diagonal space-frequency MIMO channel matrix containing the MR×MT blocks H(n) with n=1, . . . , N on its diagonal. The block-diagonal space-frequency MIMO channel matrix H possesses the desirable feature that symbols transmitted over a channel as described by H suffer only from multiple access interference which may be spatial interference, and may typically be caused by transmissions of the user himself (caused by spatial correlations within the blocks H(n)), but no longer from intersymbol interference (originally caused by the frequency-selectivity of the space-time MIMO channel). Due to the Fourier and inverse Fourier transformations applied to the space-time MIMO channel matrix {tilde over (G)} in the process of block-diagonalization, the total bandwidth of the space-time MIMO channel is considered to be divided into N subbands or sub-carriers, wherein the frequency-flat transmission channel of each of said N sub-carriers is represented by a respective MR×MT sub-carrier channel H(n), so that sub-carrier channel H(n) defines the mapping of the frequency components of transmission signals transmitted from the MT transmit antenna elements to frequency components of receive signals received at the MR receive antenna elements.
To attain that the transmitted symbols “see” only the space-frequency MIMO channel matrix H of equation (8) during their transmission, pre-processing is required at the transmitter, and post-processing is required at the receiver. Furthermore, a re-ordering of the transmit symbols zmT((n) into N MT-element transmit signal vectorsz(n)=z[z1(n) . . . zMT(n)]T with n=1, . . . , N, and a re-ordering of the receive symbols xmR(n) into N MR-element receive signal vectorsx(n)=[x1(n) . . . xMR(n)]T with n=1, . . . , N is required.
Equation (8) clearly defines the shape of the processing required at the transmitter and receiver site, which is schematically depicted in FIG. 1 (as in U.S. Pat. No. 6,144,711): At the transmitter, first a permutation operation has to be performed on the transmit signal vectors z(n) (matrix PT), and subsequently an inverse Fourier transformation is required (matrix Y(MT)H, block 101). At the receiver, first a Fourier transformation (matrix Y(MR) block 105) and then a permutation operation (matrix PR) is performed on the signals received at the MR receive antenna elements, yielding the receive signal vectors x(n). Keeping in mind that the space-time MIMO channel matrix is actually composed of Toeplitz blocks (MIMO matrix G with SISO blocks GmR,mT 103, also cf. equation (2)), and not of circulant blocks (matrix {tilde over (G)}), this difference has to be taken care of at the transmitter and receiver site. This is achieved by adding a cyclic prefix of length at least v (the last v symbols of a signal are copied and prepended before the first symbol of the respective signal) to the signals that leave the inverse Fourier transformation block 101, and by removing the first v symbols of the signals received at the MR receive antenna elements at the receiver site in a block 104.
The inverse Fourier transformation performed in blocks 101 at the transmitter site reflects the fact that the transmit symbols are considered to be frequency domain symbols that have to be transformed to the time domain in order to be transmitted over the space-time MIMO channel {tilde over (G)}. Similarly, the Fourier transformation performed in blocks 105 at the receiver site reflects that the time domain receive signals as output by the space-time MIMO channel {tilde over (G)} have to be converted back to the frequency domain.
The system set-up of FIG. 1 can then be conveniently described by a decoupled space-frequency system model:x(n)=H(n)z(n)+I(n),  (9)with n=1, . . . , N. This decoupled system model with the sub-carrier channel H(n) being represented by block 201 is depicted in FIG. 2.
The system of FIG. 2 is capable of transmitting MT transmit symbols zmT(n) over an MR×MT channel H(n), wherein the only interference arising in this transmission stems from multiple access interference between the MT transmit symbols transmitted over the same channel H(n). The space-time MIMO transmission system thus can be represented by N decoupled space-frequency transmission systems as depicted in FIG. 2.
To remove or mitigate the remaining multiple access interference between the MT transmit symbols transmitted over the same sub-carrier channel H(n), equalization can be applied to said sub-carrier channels H(n), either by applying a pre-equalization filter WT(n) at the transmitter site, a post-equalization filter WR(n) at the receiver site, or pre- and post-equalization filters WT(n) and WR(n) at both the transmitter and the receiver site.
This approach is schematically depicted in FIG. 3, wherein the pre-equalization is represented by block 301, and the post equalization is represented by block 302. At each sub-carrier channel index n=1, . . . , N, the pre-equalization filter WT(n) allows to map K symbols sk(n) with k=1, . . . , K onto the MT transmit symbols zmT(n) with mT=1, . . . , MT. Said K symbols sk(n) with n=1, . . . , N represent symbols of K signals that are to be transmitted over said transmission channel. By properly choosing the pre-equalization filter WT(n), it is not only possible to transmit a number of signals K that differs from the number of transmit antenna elements MT, but to perform pre-equalization of the sub-carrier channels H(n) in a way that, for example, multiple access interference between the K signals is mitigated or removed. In addition to or instead of said pre-equalization, a post-equalization may be performed at the receiver site, which maps the MR receive symbols xmR(n) to K reconstructed symbols ŝk(n).
For the case when only pre-equalization is performed, the post-equalization matrix may for instance equal the identity matrix, K=MR may hold, and the reconstructed symbols ŝk(n) then may equal the receive symbols H. Similarly, for the case when only post-equalization is performed, the pre-equalization matrix may equal the identity matrix, K=MT may hold, and the transmit symbols zmT(n) may equal the symbols sk(n) that are to be transmitted over the transmission channel.
For instance, if Zero-Forcing (ZF) post-equalization of the sub-carrier channels H(n) at the receiver site is desired, at each sub-carrier channel index n=1, . . . , N, the left pseudo-inverse WR(n)=(HH(n)H(n))−1HH(n) of each sub-carrier channel H(n) has to be computed based on estimates of said sub-carrier channels H(n). It is readily seen that the computation of the left pseudo inverse for each sub-carrier channel has to be performed N times and requires only MT3 complex multiplications each, so that, in contrast to ZF being applied to the space-time MIMO channel matrix G, which requires (NMT)3 complex multiplications, a huge saving in computational costs is achieved by equalizing the space-frequency channel instead of the space-time channel (the computational effort for implementing the Fourier and inverse Fourier transformations and the permutations can be neglected compared to the computational effort for the matrix inversions).
However, the remaining computational effort required to equalize the space-frequency MIMO channel can still be considerate in particular for large numbers of transmitted symbols and for large numbers of antenna elements. To reduce the computational complexity, U.S. Pat. No. 6,144,711 thus proposes to use the same equalization filters for all N sub-carrier channels H(n). With the N sub-carrier channels H(n) being roughly equal only for frequency-flat transmission channels, this proposal is however not suited for frequency-selective transmission channels, where at least some of the N sub-carrier channels H(n) are substantially different from each other.